Techniques for biasing a radio frequency digital to analog converter

ABSTRACT

Various techniques for biasing a radio frequency digital-to-analog converter are described. In one embodiment, a baseband processor may comprise a plurality of output drivers to generate a plurality of base currents for biasing a radio frequency digital-to-analog converter. The baseband processor may comprise a serial control interface to generate a programming signal for controlling a relationship among the plurality of base currents. Other embodiments are described and claimed.

RELATED APPLICATIONS

This application is related to commonly assigned U.S. patent application Ser. No. 11/398,060 titled “Baseband Signal Processor” and U.S. patent application Ser. No. 11/398,286 titled “Differential Analog Filter” filed on Apr. 4, 2006. These applications are incorporated by reference.

BACKGROUND

A mobile telecommunications transmitter, such as a polar transmitter, may include a polar digital radio frequency power amplifier (RF PA) to amplify a signal prior to transmission. In a polar transmitter, the data to be transmitted is separated into amplitude and phase signals. After separation, the phase signal is applied to a phase modulator, and the amplitude signal is applied to an amplitude modulator, such as a radio frequency digital-to-analog converter (RFDAC).

In operation, the RFDAC may be used to modulate the amplitude signal as well as to control the power that is transmitted. Accordingly, there may be a need for systems and techniques for biasing an RFDAC to improve power control and to optimize RF PA power efficiency over a range of output power.

SUMMARY

In one embodiment, a baseband processor may comprise a plurality of output drivers to generate a plurality of base currents for biasing a radio frequency digital-to-analog converter. The baseband processor may comprise a serial control interface to generate a programming signal for controlling a relationship among the plurality of base currents.

In one embodiment, a baseband processor may comprise a plurality of output drivers to generate a plurality of base currents for biasing a radio frequency digital-to-analog converter. The plurality of output drivers may receive a single power level signal to adjust the plurality of base currents.

In one embodiment, a radio frequency digital-to-analog converter may comprise a plurality of gain stages to be biased by a plurality of base currents. The plurality of base currents may be generated by a baseband processor based on a single power level signal.

In one embodiment, a baseband signal processing method may comprise generating a programming signal for controlling a relationship among a plurality of base currents and biasing a radio frequency digital-to-analog converter with the plurality of base currents to improve power efficiency.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates one embodiment of a baseband signal processing system.

FIG. 2 illustrates one embodiment of a tuning voltage generation system.

FIG. 3 illustrates one embodiment of output driver system.

FIG. 4 illustrates one embodiment of an RFDAC.

DETAILED DESCRIPTION

FIG. 1 illustrates one embodiment of a baseband signal processing system 100. As shown, the baseband signal processing system 100 may comprise an RFDAC 110 to be biased by a baseband processor 120. The RFDAC 110 may be implemented, for example, by an RF PA. In various implementations, the baseband processor 120 may be arranged to generate a plurality of base currents including for example, base current I_(B1), base current I_(B2), and base currents I_(b1)-I_(bn) for biasing the RFDAC 110.

As shown, the RFDAC 110 may comprises a plurality of gain stages including, for example, a first gain stage 111, a second gain stage 112, and a third gain stage 113 which may be AC coupled together for the carrier and phase input from the output of the second stage. In this embodiment, an RF signal Φ is input to the first gain stage 111 of the RFDAC 110, and the base current I_(B1) is applied from the baseband processor 120 to the first gain stage 111 of the RFDAC 110. As such, the first gain stage 111 is biased by the base current I_(B1) and to amplify RF signal Φ.

In the RFDAC 110, the base current I_(B2) is applied from the baseband processor 120 to the second gain stage 112. As such, the second gain stage 112 is biased by the base current I_(B2) and to further amplify RF signal Φ provided from the first gain stage 111. In general, the base current I_(B1) and the base current I_(B2) will be different because the second stage 112 input power is amplified by the first gain stage 111. Therefore, the second gain stage 112 is biased at a higher DC output point.

After being amplified by the first gain stage 111 and the second gain stage 112, the amplified RF signal Φ is sent to the third gain stage 113. In the third gain stage 113, the amplified RF signal Φ is modulated by a baseband signal and is then transmitted through the antenna 114. In this embodiment, the baseband signal may comprise an amplitude signal provided to the third gain stage 113 through the base currents I_(b1)-I_(bn). In various implementations, the combined base currents I_(b1)-I_(bn) provided from the baseband processor 120 may be used to control the amplification of the RF signal Φ provided by the third gain stage 113.

In various embodiments, the baseband processor 120 may be arranged to adjust the base currents I_(B1), I_(B2), and I_(b1)-I_(bn) to individually bias the first gain stage 111, the second gain stage 112, and the third gain stage 113 of the RFDAC 110. Since the RFDAC 110 comprises three gain stages, it may be necessary to adjust the bias for each of the gain stages individually to obtain optimal power efficiency and linearity. This may be necessary, for example, to compensate for and to minimize the effects of variations in on-chip and off-chip device parameters, such as CMOS process variations, power supply variations, temperature variations, transconductance (G_(m)) variations, and so forth. In many cases, the relationship or ratio required among the base currents applied to each gain stage of the RFDAC 110 may not be the same from chip to chip.

As shown, the RFDAC 110 may comprise a sense device 115. In various embodiments, the gain stages of the RFDAC 110 may comprise transistors implemented as hetero-junction bipolar transistor (HBT) devices (e.g., GaAs HBT devices). In such embodiments, the sense device 115 may comprise an HBT device formed integrally on the same substrate as the other transistors of the RFDAC 110 so that the variations due to process variables for the other transistors of the RFDAC 110 and the sense device 115 are similar.

In various implementations, the baseband processor 120 may be arranged to detect the ratio of the collector-emitter current to the base-emitter current, or current gain (β), of the sense device 115. From the detected β of the sense device 115, the baseband processor 120 may generate a compensation signal that is proportional to 1/β. In various embodiments, the 1/β compensation signal may be used by the baseband processor 120 to generate the base currents I_(B1), I_(B2), and I_(b1)-I_(bn).

In one embodiment, the baseband processor 120 may output a current MV_B to the base of the sense device 115 (e.g., HBT device) and may provide a fixed precision current MV_C to the collector of the sense device 115 to determine the β of the sense device 115. The baseband processor 120 may automatically adjust the current MV_B by sensing and maintaining the voltage at the collector of the sense device 115 such that the collector voltage will be high enough to maintain the sense device 115 in its linear operating range. When such condition is achieved, through the adjustment of the current MV_B, the resulting current MV_B is 1/β of the current MV_C.

In various embodiments, the baseband processor 120 may be implemented by an analog baseband signal processor and/or an amplitude baseband processing chip, for example. As shown, the baseband processor 120 may include a power control portion 130, a filter portion 140, and a driver portion 150. The power control portion 130, the filter portion 140, and the driver portion 150 are described in further detail below as well as in commonly assigned U.S. patent application Ser. No. 11/398,060 titled “Baseband Signal Processor” and U.S. patent application Ser. No. 11/398,286 titled “Differential Analog Filter,” which are incorporated by reference.

The power control portion 130 may comprise a serial control interface 131. The serial control interface 131 may be arranged to provide a communication link from a computer, such as a personal computer (PC) to the baseband processor 120. In various implementations, the serial control interface 131 may provide a user with the ability to access and/or program various elements within the baseband processor 120. For example, the serial control interface 121 may enable programming of elements within the driver portion 150.

In one embodiment, the serial control interface 131 may be arranged to generate a programming signal for controlling the set of base currents I_(B1), I_(B2), and I_(b1)-I_(bn) provided to the RFDAC 110. For example, the serial control interface 131 may allow a user to send a programming signal to the driver portion 150 of the RFDAC 110 for individually adjusting the base currents I_(B1), I_(B2), and I_(b1)-I_(bn).

In various implementations, the serial control interface 131 may enable a user to maintain a fixed ratio among the output base currents I_(B1), I_(B2), and I_(b1)-I_(bn). The ratio among the base current I_(B1), the base current I_(B2), and the base currents I_(b1)-I_(n) may be controlled, for example, to improve and/or optimize RF PA power efficiency. As such, RF PA power efficiency may be maintained over a broad range of output power. In some embodiments, further improvement and/or optimization may be achieved by additional individual fine-tuning of the base currents I_(B1), I_(B2), and I_(b1)-I_(bn).

In various embodiments, the serial control interface 131 may be used to adjust Gm values of the driver portion 150 for biasing individual stages of the RFDAC 110. The Gm values of the driver portion 150 may be adjusted to control the base currents I_(B1), I_(B2), and I_(b1)-I_(bn) in order to improve and/or optimize RF PA efficiency. In various implementations, once the optimal biasing conditions and/or Gm values are determined, the setting may be latched into the memory of the serial control interface 131.

In various embodiments, the baseband processor 120 may be arranged to provide unified or “single point” power control for establishing the output power of an RF PA. In one embodiment, for example, the baseband processor 120 may be arranged to use a single power level signal to adjust the base currents I_(B1), I_(B2), and I_(b1)-I_(bn) for all three stages of the RFDAC 110 simultaneously. In various implementations, the power level signal may comprise a single power control voltage applied through the filter portion 140 to the driver portion 150 creating a central or “one knob” control point.

Referring to the embodiment of FIG. 1, for example, the “one knob” control signal may refer to the input signal “POWER LEVEL SIGNAL.” In this embodiment, two sets of power control signals may be generated from this input signal. The first set of power control signals may include signals that are DC in nature but proportional to the input POWER LEVEL SIGNAL. The first set of power control signals may be applied to the driver portion 150 for the first gain stage 111 and the second gain stage 112 of the RFDAC 110.

The second set of power control signals may comprise n (e.g., n=11) analog signals swinging between two levels and having amplitude proportional to the input POWER LEVEL SIGNAL. In this embodiment, the source of these n analog signals may comprise n 1-bit DACs or analog multiplexers structured and arranged so that their analog outputs are proportional to the POWER LEVEL SIGNAL. Tracing back a step further, the source of the n analog signals may comprise n digital “AMPLITUE SIGNALS” as shown in FIG. 1.

In various embodiments, when this central power control signal is varied, all output currents, I_(B1), I_(B2) and I_(b1) . . . I_(bn) will change simultaneously proportional to constants G_(m1), G_(m2), and G_(m3) respectively giving the effect of “one knob” control to the output power at RFDAC 110. The proportionality constants G_(m1), G_(m2), and G_(m3), may be individually programmable through the serial control interface 131 and registers residing there. When applied to the driver portion 150, the single power control voltage may bias multiple stage of the RFDAC 110 simultaneously based on the G_(m) values and establish the output power of the RF PA.

The baseband processor 120 may be arranged to receive inputs including, for example, amplitude signals and the power level signal. In some cases, the inputs to the baseband processor 120 may be received from a digital signal processor and/or baseband integrated circuit implementing a coordinate rotation digital computer (CORDIC) algorithm.

In various implementations, the baseband processor 120 may receive input data separated into amplitude and phase signals. For example, the RFDAC 110 may be used to modulate an input in-phase/quad-phase (IQ) baseband signal. Before the IQ baseband signal is applied to the RFDAC 110, it is first divided into amplitude and phase signals. The amplitude signals may be quantized by the baseband processor 120 and output to the RFDAC 110, whose RF input is separately modulated by the phase component. In some embodiments, the RFDCA 110 may comprise the last stage of a polar transmitter, making it possible to reduce current drain quickly as the transmit power level is reduced.

As shown, the power control portion 130 may include one or more analog multiplexer and level shift modules 132-1-x, where x represents any positive integer value. In various implementations, the analog multiplexer and level shift modules 132-1-x may be arranged to receive amplitude signals and the power level signal. The amplitude signals may comprise, for example, one or more digital amplitude baseband signals. In one embodiment the amplitude signal may comprise an n-bit (e.g., 11-bit) coded digital amplitude signal.

The analog multiplexer and level shift modules 132-1-x may be arranged to convert the n-bit coded digital amplitude signal to analog. In various embodiments, the analog multiplexer and level shift blocks 132-1-x may perform digital-to-analog conversion (DAC) according to the input power level resulting in amplitude scaled to the power level. For example, a code of 1 input to the analog multiplexer and level shift modules 132-1-x may be converted to 200 mV, referenced to a common mode DC level, according to the power level.

The baseband processor 120 may comprise a filter portion 140 including a plurality of filters such as a first filter 141 and a bank of filters 142-1-y, where y represents any positive integer value. In various embodiments, the first filter 141 and the bank of filters 142-1-y may be arranged in parallel to filter the output from the analog multiplexer and level shift modules 132-1-x.

The filter portion 140 may employ various types of filters, such as low-pass filters (LPFs) having a predetermined cut-off frequency. In some cases, each LPF may be implemented as a linear filter, such as a Bessel filter with maximally flat group delay (linear phase response) and small overshoot. For example, the LPFs may be implemented as third-order Bessel filters. In low power consumption embodiments, the filter portion 140 may employ a Sallen-Key architecture cascaded by a passive RC network. The embodiments are not limited in this context.

The baseband processor 120 may comprise a driver portion 150 including a plurality of tuning voltage (V_(tune)) generators such as a V_(tune1) generator 151, a V_(tune2) generator 152, and a V^(tune3) generator 153. While three V_(tune) generators are shown by way of example, it can be appreciated that a greater or fewer number of V_(tune) generators may be used in other implementations. In one embodiment, the V_(tune1) generator may generate a tuning voltage V_(tune1), the V_(tune2) generator may generate a tuning voltage V_(tune2), and the V^(tune3) generator may generate a tuning voltage V_(tune3). As shown, the tuning voltage V_(tune3) may be amplified or buffered by an amplifier 154, such as a unity-gain amplifier.

In various implementations, the V_(tune) generators may be independently controlled by the serial control interface 131. Each of the V_(tune) generators may comprise, for example, a current multiplier (I-multiplier or current-multiplying-DAC) configured to be independently programmed by the serial control interface 131. By independently controlling each of the V_(tune) generators, the serial control interface 131 may be arranged to control the ratio of the base currents I_(B1), I_(B2), and I_(b1)-I_(bn) provided to each stage of the RFDAC 110.

In one embodiment, the V_(tune1) generator 151, the V_(tune2) generator 152, and the V_(tune3) generator 153 each may comprise a 5-bit programmable I-multiplier, and the serial control interface 131 may generate a 15-bit programming signal. The 15-bit programming signal from the serial control interface 131 may be divided into three 5-bit groups, with each 5-bit group independently programming a separate I-multiplier. Using the 5-bits from the serial control interface 131, the multiplication factor for each I-multiplier of the V_(tune1) generator 151, the V_(tune2) generator 152, and the V_(tune3) generator 153 may be programmed.

In various embodiments, the tuning voltages V_(tune1), V_(tune2), and V_(tune3) may be different but proportional with respect to the values of the three 5-bit digital values, respectively. In one embodiment, the V_(tune) generators may be arranged to generate tuning voltages V_(tune1), V_(tune2), and V_(tune3) in proportion of 1:2:3 with respect to each other. For example, the V_(tune1) generator may comprise an I-multiplier programmed with a multiplication factor (K₁) equal to one, the V_(tune2) generator may comprise an I-multiplier programmed with a multiplication factor (K₂) equal to two, and the V_(tune3) generator may comprise an I-multiplier programmed with a multiplication factor (K₃) equal to three. After being programmed, the I-multipliers of the V_(tune) generators may be arranged to multiply a current I_(D) individually by K₁, K₂, and K₃ to generate currents I_(D1), I_(D2), and I_(D3), which are used to generate the tuning voltages V_(tune1), V_(tune2), and V_(tune3).

As shown, the driver portion 150 may comprise a plurality of output drivers including a G_(m1) driver 155, a G_(m2) driver 156, and G_(m3) drivers 157-1-n, where n represents any positive integer value. In various embodiments, the tuning voltages V_(tune1), V_(tune2), and V_(tune3) may be provided to the output drivers including the G_(m1) driver 155, the G_(m2) driver 156, and the G_(m3) drivers 157-1-n. Collectively, the tuning voltages V_(tune1), V_(tune2), and V_(tune3) may be used to adjust and control the G_(m) value of each of the output drivers to provide optimized RF PA power efficiency over a range of output power.

In one embodiment, the tuning voltage V_(tune1) may be provided to the G_(m1) driver 155, the tuning voltage V_(tune2) may be provided to the G_(m2) driver 156, and the tuning voltage V_(tune3) may be provided to the G_(m3) drivers 157-1-n. Each of the G_(m1) driver 155, the G_(m2) driver 156, and the G_(m3) drivers 157-1-n may comprise, for example, a multiplier for multiplying the tuning voltages V_(tune1), V_(tune2), and V_(tune3). In various implementations, the values G_(m1), G_(m2), and G_(m3) may be adjusted to provide a constant ratio among the G_(m1) driver 155, the G_(m2) driver 156, and the G_(m3) drivers 157-1-n. By adjusting the values G_(m1), G_(m2), and G_(m3) the amount by which the base currents I_(B1), I_(B2), and I_(b1)-I_(bn) are adjusted may be controlled. For example, the ratio among the base current I_(B1) provided by the G_(m1) driver 155, the base current I_(B2) provided by the G_(m2) driver 156, and the base currents I_(b1)-I_(n) provided by the G_(m3) drivers 157-1-n may be controlled to allow optimization of RF PA power efficiency in the RFDAC 110.

In various embodiments, the G_(m1) driver 155, the G_(m2) driver 156, and the Gm₃ drivers 157-1-n may be arranged to use a single power level signal to simultaneously adjust the base currents I_(B1), I_(B2), and I_(b1)-I_(bn). In such embodiments, the power level may be filtered by the first filter 141 and sent to the G_(m1) driver 155 and to the G_(m2) driver 156. The power level also may be filtered by the bank of filters 142-1-n and sent to the G_(m3) drivers 157-1-n. The power level may comprise, for example, a single power control voltage applied to the G_(m1) driver 155, the G_(m2) driver 156, and the G_(m3) drivers 157-1-n.

In various implementations, all of the output drivers including the G_(m1) driver 155, the G_(m2) driver 156, and the G_(m3) drivers 157-1-n maybe adjusted by a certain amount according to the power level. As such, a single power level signal may be used to bias multiple stage of the RFDAC 110 simultaneously based on the G_(m) values. In various implementations, the power level signal may comprise a single power control voltage applied to the driver portion 150 creating a central or “one knob” control point. In such implementations, unified or “single point” power control is provided for establishing the output power of an RF PA.

Referring to the embodiment of FIG. 1, for example, the “one knob” control signal may refer to the input POWER LEVEL SIGNAL. In this embodiment, two sets of power control signals are generated from the POWER LEVEL SIGNAL. The first set of power control signals includes signals that are DC in nature but proportional to the input POWER LEVEL SIGNAL. The first set of power control signals are applied to the G_(m1) driver 155 and the G_(m2) driver 156 for the first gain stage 111 and the second gain stage 112 of the RFDAC 110.

The second set of power control signals may comprise n (e.g., n=11) analog signals swinging between two levels and having amplitude proportional to the input POWER LEVEL SIGNAL. In this embodiment, the source of these n analog signals may comprise 1-bit DACs, such as analog multiplexer and level shift blocks 132-1-x shown in FIG. 1. The analog multiplexer and level shift blocks 132-1-x may be structured and arranged so that their analog outputs are proportional to the POWER LEVEL SIGNAL. Tracing back a step further, the source of the n analog signals may comprise n digital AMPLITUE SIGNALS as shown in FIG. 1.

In this embodiment, when the central power control signal is varied, all output currents, I_(B1), I_(B2) and I_(b1) . . . I_(bn) will change simultaneously proportional to the constants G_(m1), G_(m2), and G_(m3) respectively giving the effect of “one knob” control to the output power at RFDAC 110. The proportionality constants G_(m1), G_(m2), and G_(m3) are individually programmable through the serial control interface 131 and registers residing there. When applied to the G_(m1) driver 155, the G_(m2) driver 156, and the G_(m3) drivers 157-1-n, the single power control voltage may bias multiple stage of the RFDAC 110 simultaneously based on the G_(m) values and establish the output power of the RF PA.

In various embodiments, the output drivers including the G_(m1) driver 155, the G_(m2) driver 156, and the G_(m3) drivers 157-1-n may be biased to compensate for variations in on-chip and off-chip device parameters, such as CMOS process variations, power supply variations, temperature variations, G_(m) variations, and so forth.

As shown, the driver portion 150 may comprise a 1/β generator 158. In various implementations, the 1/βgenerator 158 may be arranged to detect the β of the sense device 115 of the RFDAC 110. From the detected β of the sense device 115, the 1/β generator 158 may generate a compensation signal that is proportional to 1/β.

In various embodiments, the output drivers including the G_(m1) driver 155, the G_(m2) driver 156, and the G_(m3) drivers 157-1-n may generate base currents I_(B1), I_(B2), and I_(b1)-I_(bn) based on the signal proportional to 1/β. In one embodiment, for example, the 1/β generator 158 may provide the 1/β compensation signal to the V_(tune1) generator 151, the V_(tune2) generator 152, and the V_(tune3) generator 153. When biased by the tuning voltages V_(tune1), V_(tune2), and V_(tune3), the G_(m1) driver 155, the G_(m2) driver 156, and the Gm₃ drivers 157-1-n may be arranged to generate base currents I_(B1), I_(B2), and I_(b1)-I_(bn) which are compensated for β variation in the RFDAC 110. Base currents (e.g., I_(B1), I_(B2), and I_(b1)-I_(bn)) carrying the characteristic of 1/β may be multiplied by β so that the collector currents of devices in the RFDAC 110 are substantially independent of variations in β.

FIG. 2 illustrates one embodiment of a tuning voltage generation system 200. In various embodiments, the tuning voltage generation system 200 may be implemented by the baseband processor 120 depicted in FIG. 1. The embodiments, however, are not limited in this context.

As shown, the tuning voltage generation system 200 comprises a serial control interface 210, a 1/β generator 220, and a plurality of V_(tune) generators including a V_(tune1) generator 230, a V_(tune2) generator 240, and a V^(tune3) generator 250. A common supply voltage V_(DD) (e.g., 2.85V) is applied to the 1/β generator 220, the V_(tune1) generator 230, the V_(tune2) generator 240, and the V_(tune3) generator 250. While three V_(tune) generators are shown by way of example, it can be appreciated that a greater or fewer number of V_(tune) generators may be used in other implementations.

The 1/β generator 220 comprises a PMOS device 221 (e.g., P-MOSFET) coupled to an HBT device 222 (e.g., GaAs HBT device). In one embodiment, the β varies from 40 to 140 and ηβ≈52 and τβ≈15% for the HBT device 222. The HBT device 222 is coupled to a current source 223 and an amplifier 224. The current source 223 generates a reference current I_(ref) (e.g., 20 μA) which is fed into the collector of the HBT device 222, and the amplifier 224 drives the gate of the PMOS device 221.

A common mode voltage V_(cm) (e.g., 1.1V) is coupled to the non-inverting (+) input of the amplifier 224. Because little or no current flows into the inverting (−) and non-inverting (+) inputs of the amplifier 224, the input voltages at the inverting (−) and non-inverting (+) inputs are substantially equal, and the voltage at the inverting (−) input of the amplifier 224 and the collector of the HBT device 222 is V_(cm). If the collector of the HBT device 222 is kept at the voltage level V_(cm), the HBT device 222 is guaranteed to operate in the linear region. If not at V_(cm), the amplifier 224 will generate a compensating voltage such as a negative pulling voltage if the voltage is too high.

The V_(tune1) generator 230 comprises a transistor 231. In various embodiments, the transistor 231 may be similar to and closely match the PMOS device 221. The source of the transistor 231 is coupled to the common supply voltage V_(DD), and the gate of the transistor 231 is driven by the output of the amplifier 224. Accordingly, the current I_(D) driven by the transistor 231 is proportional to 1/βand is equal to I_(ref)/β. In the V_(tune1) generator 230, the drain current I_(D) is multiplied by a multiplier K₁ 232 to generate a current I_(D1). The multiplier K₁ 232 may be individually controlled by the serial control interface 210.

The V_(tune1) generator 230 comprises an amplifier 233. The output of the amplifier 233 is coupled to the gate of a transistor M_(1c) 234, and the inverting (−) input of the amplifier 233 is coupled to the drain of the transistor M_(1c) 234. The common mode voltage V_(cm) is coupled to the non-inverting (+) input of the amplifier 233 and is coupled to the gate of a transistor M₁ 235. Accordingly, the voltage at the inverting (−) input of the amplifier 233 and the drain of the transistor M_(1c) 234 is V_(cm).

The source of the transistor M_(1c) 234 is coupled to the drain of the transistor M₁ 235 at a node 236. The current I_(D)I is equal to the drain current in the transistor M_(1c) 234 and is forced into the transistor M₁ 235 to generate the bias control signal V_(tune1) at the node 236. The tuning voltage V_(tune1) may be a voltage that keeps the transistor M₁ 235 in a saturated or final state. In one embodiment, the transistors M_(1c) 234 and M₁ 235 may comprise N-MOSFET transistors biased in triode mode, for example.

The V_(tune2) generator 240 comprises a transistor 241. In various embodiments, the transistor 241 may be similar to and closely match the PMOS device 221. The source of the transistor 241 is coupled to the common supply voltage V_(DD), and the gate of the transistor 241 is driven by the output of the amplifier 224. Accordingly, the current I_(D) driven by the transistor 241 is proportional to 1/β and is equal to I_(ref)/β. In the V_(tune2) generator 240, the drain current I_(D) is multiplied by a multiplier K₂ 242 to generate a current I_(D2). The multiplier K₂ 242 may be individually controlled by the serial control interface 210.

The V_(tune2) generator 240 comprises an amplifier 243. The output of the amplifier 243 is coupled to the gate of a transistor M₂c 244, and the inverting (−) input of the amplifier 243 is coupled to the drain of the transistor M₂c 244. The common mode voltage V_(cm) is coupled to the non-inverting (+) input of the amplifier 243 and is coupled to the gate of a transistor M₂ 245. Accordingly, the voltage at the inverting (−) input of the amplifier 243 and the drain of the transistor M_(2c) 244 is V_(cm).

The source of the transistor M_(2c) 244 is coupled to the drain of the transistor M₂ 245 at a node 246. The current I_(D2) is equal to the drain current in the transistor M_(2c) 244 and is forced into the transistor M₂ 245 to generate the bias control signal V_(tune2) at the node 246. The tuning voltage V_(tune2) may be a voltage that keeps the transistor M₂ 245 in a saturated or final state. In one embodiment, the transistors M_(2c) 244 and M₂ 245 may comprise N-MOSFET transistors biased in triode mode, for example.

The V_(tune3) generator 250 comprises a transistor 251. In various embodiments, the transistor 251 may be similar to and closely match the PMOS device 221. The source of the transistor 251 is coupled to the common supply voltage V_(DD), and the gate of the transistor 251 is driven by the output of the amplifier 224. Accordingly, the current I_(D) driven by the transistor 251 is proportional to 1/β and is equal to I_(ref)/β. In the V_(tune3) generator 250, the drain current I_(D) is multiplied by a multiplier K₃ 252 to generate a current I_(D3). The multiplier K₃ 252 may be individually controlled by the serial control interface 210.

The V_(tune3) generator 250 comprises an amplifier 253. The output of the amplifier 253 is coupled to the gate of a transistor M_(3c) 254, and the inverting (−) input of the amplifier 253 is coupled to the drain of the transistor M_(3c) 254. The common mode voltage V_(cm) is coupled to the non-inverting (+) input of the amplifier 253 and is coupled to the gate of a transistor M₃ 255. Accordingly, the voltage at the inverting (−) input of the amplifier 253 and the drain of the transistor M_(3c) 254 is V_(cm).

The source of the transistor M_(3c) 254 is coupled to the drain of the transistor M₃ 255 at a node 256. The current I_(D3) is equal to the drain current in the transistor M_(3c) 254 and is forced into the transistor M₃ 255 to generate the bias control signal V_(tune3) at the node 256 through an amplifier 257. The tuning voltage V_(tune3) may be a voltage that keeps the transistor M₃ 255 in a saturated or final state. In one embodiment, the transistors M_(3c) 254 and M₃ 255 may comprise N-MOSFET transistors biased in triode mode, for example.

FIG. 3 illustrates one embodiment of an output driver system 300. In various embodiments, the output driver system 300 may be implemented by the baseband processor 120 depicted in FIG. 1. The embodiments, however, are not limited in this context.

The output driver system 300 may comprise a plurality of output drivers including G_(m3) drivers 310-1-n, where n may be equal to any positive integer value (e.g., n=11). As shown, some of the output drivers such as output drivers 310-1-4 (e.g., ×1, ×2, ×4, and ×8) may be binary weighted, and some of the output drivers 310-5-n may be weighted equally (e.g., ×N). In one embodiment, for example, ×N may be equal to ×16. The embodiments, however, are not limited in this context.

In various embodiments, the tuning voltage V_(tune3) may be provided to the G_(m3) drivers 310-1-n to adjust and control the Gm₃ value to provide optimized RF PA power efficiency over a range of output power. Each of the G_(m3) drivers 310-1-n may comprise, for example, a multiplier for multiplying the tuning voltage V_(tune3). In various implementations, the value G_(m3) may be adjusted to provide a constant ratio among the G_(m3) output drivers 310-1-n and other output drivers (e.g., a G_(m1) driver and a G_(m2) driver). By adjusting the value G_(m3), the amount by which the base currents I_(b1)-I_(bN) _(—) _(n) are adjusted may be controlled.

As shown, the G_(m3) drivers 310-1-n may be adjusted by a certain amount according to the power level. In various embodiments, the G_(m3) drivers 310-1-n may be arranged to use a single power level signal to simultaneously adjust the base currents I_(b1)-I_(bN) _(—) _(n). In such embodiments, the power level may be filtered by a bank of filters, for example, and sent to the G_(m3) drivers 310-1-n.

In some implementations, the power level may comprise, a single power control voltage applied to the G_(m3) drivers 310-1-n as well as to other drivers (e.g., a G_(m1) driver and a G_(m2) driver). In such implementations, unified or “single point” power control is provided for establishing the output power of an RF PA.

In various embodiments, the G_(m3) drivers 310-1-n may be biased to compensate for variations in on-chip and off-chip device parameters, such as CMOS process variations, power supply variations, temperature variations, G_(m) variations, and so forth. For example, the G_(m3) drivers 310-1-n may generate base currents I_(b1)-I_(bN) _(—) _(n) based on a signal proportional to 1/β.

In effect, the combined base currents I_(b1)-I_(bN) _(—) _(n) form an amplitude signal comprising an n-bit digital word (e.g., 11-bit digital word). In various embodiments, each bit of the n-bit digital word may correspond to a switching transistor in an RFDAC gain stage. Each switching transistor may be biased ON or OFF depending on the corresponding bit value. By biasing certain switching transistors ON and OFF, the combined base currents I_(b1)-I_(bN) _(—) _(n) may be used to effectively control the amplification provided by the RFDAC gain stage.

FIG. 4 illustrates one embodiment of an RFDAC 400. In various embodiments, the RFDAC 400 may comprise or be implemented as the RFDAC 110 depicted in FIG. 1. The embodiments, however, are not limited in this context.

As shown, the RFDAC 400 may comprise a plurality of gain stages including a first gain stage 410, a second gain stage 420, and a third gain stage 430 which may be AC coupled together. In this embodiment, the first gain stage 410 comprises a coupling capacitor 411 and a transistor 412 having an inductor 413 as a load. The second gain stage 420 comprises a coupling 421 capacitor and a transistor 422 having an inductor 423 as a load. The inductors 413, 423 may be used to amplify AC but not DC signals.

In this embodiment, the third gain stage 430 comprises a coupling capacitor 431 and a plurality of switching transistors 432-1-n, where n may represent any positive integer value (e.g., n=11). The third gain stage 430 also comprises a plurality of resistors 433-1-n and a plurality of capacitors 434-1-n coupled to the bases of the switching transistors 432-1-n. As shown, some transistors such as switching transistors 432-1-4 (e.g., ×1, ×2, ×4, and ×8) may be binary weighted, and some transistors such as switching transistors 432-5-n may be weighted equally (e.g., ×N). In one embodiment, for example, ×N may be equal to ×16. The embodiments, however, are not limited in this context.

In various implementations, an RF signal Φ and a base current I_(B1) are applied to the first gain stage 410 of the RFDAC 200. In the first gain stage 410, the input RF signal Φ is applied to the capacitor 411, and the base current I_(B1) is applied to the base of the transistor 412. As such, the transistor 412 is biased by the base current I_(B1) and is modulated by the RF signal Φ.

In the second gain stage 420, a base current I_(B2) is applied to the base of transistor 422. The transistor 422 is biased by the base current I_(B2) and is modulated by the RF signal that was amplified by the first gain stage 410. In general, the base current I_(B1) and the base current I_(B2) will be different because the second gain stage 420 is amplified by the first gain stage 410. Therefore, the second gain stage 420 is biased at a higher DC output point.

After being amplified by the first gain stage 410 and the second gain stage 420, the amplified RF signal Φ is sent to the third gain stage 430. In the third gain stage 430, the amplified RF signal Φ is modulated by a baseband signal and is then transmitted through the antenna 435. In this embodiment, the baseband signal may comprise an amplitude signal provided to the third gain stage 430 through base currents I_(b1)-I_(bN) _(—) _(n).

In effect, the combined base currents I_(b1)-I_(bN) _(—n) form an amplitude signal comprising an n-bit digital word (e.g., 11-bit digital word). Each bit of the n-bit digital word corresponds to one of the switching transistors 432-1-n, and each one of the switching transistors 432-1-n is biased ON or OFF depending on the corresponding bit value. By biasing certain switching transistors ON and OFF, the amplification of the RF signal Φ provided by the third gain stage 430 may be effectively controlled.

As shown, the RFDAC 400 may comprise a sense device 436 implemented an HBT device formed integrally on the same substrate as the other transistors of the RFDAC 400 so that the variations due to process variables for the other transistors of the RFDAC 400 and the sense device 436 are similar. In various implementations, a current MV_B may be received at the base of the sense device 436 and a fixed precision current MV_C may be received at the collector of the sense device 436 for determining the β of the sense device 436.

Numerous specific details have been set forth herein to provide a thorough understanding of the embodiments. It will be understood by those skilled in the art, however, that the embodiments may be practiced without these specific details. In other instances, well-known operations, components and circuits have not been described in detail so as not to obscure the embodiments. It can be appreciated that the specific structural and functional details disclosed herein may be representative and do not necessarily limit the scope of the embodiments.

It is also worthy to note that any reference to “one embodiment” or “an embodiment” means that a particular feature, structure, or characteristic described in connection with the embodiment is included in at least one embodiment. The appearances of the phrase “in one embodiment” in various places in the specification are not necessarily all referring to the same embodiment.

Some embodiments may be implemented using an architecture that may vary in accordance with any number of factors, such as desired computational rate, power levels, heat tolerances, processing cycle budget, input data rates, output data rates, memory resources, data bus speeds and other performance constraints. The embodiments are not limited in this context.

While certain features of the embodiments have been illustrated as described herein, many modifications, substitutions, changes and equivalents will now occur to those skilled in the art. It is therefore to be understood that the appended claims are intended to cover all such modifications and changes as fall within the true spirit of the embodiments. 

1. A baseband processor comprising: a plurality of output drivers to generate a plurality of base currents for biasing a radio frequency digital-to-analog converter; a serial control interface to generate a programming signal for controlling a relationship among said plurality of base currents; and a plurality of tuning voltage generators to generate a plurality of tuning voltages for biasing said plurality of output drivers, each of said plurality of tuning voltage generators comprising a current multiplier to be programmed by said serial control interface.
 2. The baseband processor of claim 1, said relationship comprising a fixed ratio among said plurality of base currents.
 3. A baseband processor comprising: a plurality of output drivers to generate a plurality of base currents for biasing a radio frequency digital-to-analog converter; a serial control interface to generate a programming signal for controlling a relationship among said plurality of base currents; and a plurality of tuning voltage generators to generate a plurality of tuning voltages for biasing said plurality of output drivers said serial control interface to control a ratio among said plurality of tuning voltages.
 4. A baseband processor comprising: a plurality of output drivers to generate a plurality of base currents for biasing a radio frequency digital-to-analog converter; a serial control interface to generate a programming signal for controlling a relationship among said plurality of base currents; and a plurality of tuning voltage generators to generate a plurality of tuning voltages for biasing said plurality of output drivers, said plurality of tuning voltage generators to adjust transconductance values of said plurality of output drivers.
 5. The baseband processor of claim 1, said plurality of output drivers to receive a single power level signal to adjust said plurality of base currents.
 6. The baseband processor of claim 5, said power level signal comprising a single power control voltage.
 7. A baseband processor comprising: a plurality of output drivers to generate a plurality of base currents for biasing a radio frequency digital-to-analog converter, said plurality of output drivers to receive a single power level signal to adjust said plurality of base currents; a serial control interface to generate a programming signal for controlling a relationship among said plurality of base currents; and one or more analog multiplexers to receive said single power level signal.
 8. The baseband processor of claim 7, said one or more analog multiplexers to receive an amplitude signal and to scale said amplitude signal according to said single power level signal.
 9. A baseband processor comprising: a plurality of output drivers to generate a plurality of base currents for biasing a radio frequency digital-to-analog converter said plurality of output drivers to receive a single power level signal to adjust said plurality of base currents; a serial control interface to generate a programming signal for controlling a relationship among said plurality of base currents; and one or more filters to provide said single power level signal to said plurality of output drivers.
 10. A baseband processor comprising: a plurality of output drivers to generate a plurality of base currents for biasing a radio frequency digital-to-analog converter; and a serial control interface to generate a programming signal for controlling a relationship among said plurality of base currents, said baseband processor to detect a current gain (β) of said radio frequency digital-to-analog converter and to generate a compensation signal proportional to 1/β.
 11. The baseband processor of claim 10, further comprising a plurality of voltage tuning generators to generate a plurality of tuning voltages based on said compensation signal.
 12. The baseband processor of claim 1, said radio frequency digital-to-analog converter comprising a plurality of gain stages to be biased by said plurality of base currents.
 13. The baseband processor of claim 3, said plurality of output drivers to receive a single power level signal to adjust said plurality of base currents.
 14. The baseband processor of claim 13, said power level signal comprising a single power control voltage.
 15. The baseband processor of claim 3, said radio frequency digital-to-analog converter comprising a plurality of gain stages to be biased by said plurality of base currents.
 16. The baseband processor of claim 4, said plurality of output drivers to receive a single power level signal to adjust said plurality of base currents.
 17. The baseband processor of claim 16, said power level signal comprising a single power control voltage.
 18. The baseband processor of claim 4, said radio frequency digital-to-analog converter comprising a plurality of gain stages to be biased by said plurality of base currents.
 19. The baseband processor of claim 7, said power level signal comprising a single power control voltage.
 20. The baseband processor of claim 7, said radio frequency digital-to-analog converter comprising a plurality of gain stages to be biased by said plurality of base currents.
 21. The baseband processor of claim 9, said power level signal comprising a single power control voltage.
 22. The baseband processor of claim 9, said radio frequency digital-to-analog converter comprising a plurality of gain stages to be biased by said plurality of base currents. 